This invention relates to a radio communication apparatus and, more particularly, to a radio communication apparatus in which carriers are divided into two systems, namely high- and low-frequency systems, and which has a transmitting unit in each system that executes processing for transmitting a signal wirelessly, and a distortion compensating unit in each system that compensates for distortion produced by the transmitting unit.
As the field of mobile communications systems has progressed, the quantity of signals that the radio communication apparatus is required to transceive has increased. With a multicarrier-type radio communication apparatus that transmits a number of adjacent carriers, the general approach is to increase the number of carriers to deal with an increase in quantity of transceive signals.
FIG. 5 is a diagram illustrating the structure of a first multicarrier-type radio communication apparatus according to the prior art, and FIG. 6 is a diagram useful in describing the operation of this apparatus, in which (1) to (7) illustrate spectrums associated with portions identified by the same numbers in FIG. 5.
A baseband processor (BB) 1a in each of signal processing units 11 to 1n provided for respective ones of the carriers subjects data to be transmitted to baseband signal processing such as appending of error-correcting/detecting code, interleaving, multivalued modulation and code spreading and outputs a complex baseband signal. Root Nyquist filters 1b, 1c apply root Nyquist filter processing to the real and imaginary parts of the complex baseband signal that enters from the baseband processor 1a [see (1) in FIG. 6], and carrier shifters 1d, 1e multiply the real and imaginary parts of the complex baseband signal by exp(jωjt) (=1 to n) to obtain and output a carrier signal of angular frequency (ωj [see (2) in FIG. 6].
Combiners 2a, 2b perform combining for every in-phase part and quadrature-phase part of each carrier and output an in-phase component Ich and quadrature-phase component Qch of the combined-carrier signal. A distortion compensating unit (APD, or Adaptive Predistortion Device) 3 generates a distortion compensating signal that compensates for distortion produced by a transmit amplifier (described later). Arithmetic units 4a, 4b add the distortion compensating signal to the in-phase component Ich and quadrature-phase component Qch of the combined-carrier signal and inputs the resultant signals to a quadrature modulator 5. The latter applies quadrature modulation to the combined-carrier signal using a local signal the frequency of which is 1.5×W0, where W0 represents the frequency bandwidth of the combined-carrier signal [see (3) in FIG. 6]. Owing to such quadrature modulation, each carrier frequency becomes an intermediate-frequency signal raised by 1.5×W0.
A DA converter 6 converts the signal that is output from the quadrature modulator 5 to an analog signal at a sampling frequency Fs, a low-pass filter 7 eliminates unwanted high-frequency components [see (4) in FIG. 6], and a mixer 8 multiplies the input signal by a local signal that is output from a local oscillator 9, thereby up-converting the input signal to a radio signal having a frequency (f0+1.5×W0). A transmit amplifier 10 amplifies the radio signal whose frequency has been up-converted and transmits the amplified signal via an antenna [see (5) in FIG. 6].
The transmit amplifier 10 produces distortion D1, D2 on both sides of a main signal, as indicated at (5) in FIG. 6. In order to compensate for the distortion D1, D2, a directional coupler 11 extracts part of the transmit signal, a mixer 12 multiplies the input signal by the local signal that is output from the local oscillator 9, thereby down-converting the input signal to an intermediate-frequency signal of frequency 1.5×W0, and an AD converter 13 samples the input signal at a frequency Fs of 6×W0 and converts the signal to a digital signal [see (6) in FIG. 6]. A higher-order spectrum is generated (an image appears) by the AD conversion at the sampling frequency Fs. A quadrature demodulator 14 applies quadrature demodulation to the output signal of the AD converter using the local signal of frequency 1.5×W [see (7) in FIG. 6] and inputs the demodulated signal to the distortion compensating unit 3.
The distortion compensating unit 3 generates a distortion compensating signal for removing distortion in the transmit amplifier 10 using an adaptive algorithm, appends the distortion compensating signal to the transmit signal and eliminates the distortion D1, D2 on both sides of the main signal [see (5) in FIG. 6]. FIG. 7 illustrates an example of the structure of the distortion compensating unit 3, which is an adaptive predistortion device. A multiplier 3a multiplies a transmit signal x(t), which is output from the combiners 2a, 2b, by a distortion compensation coefficient hn−1(p), and an arithmetic unit 4 subtracts a distortion compensating signal hn−1·s(t) from the transmit signal x(t), thereby outputting power p [=hn(p)x(t)−x(t)] of the transmit signal x(t) after compensation. An arithmetic unit 3b calculates power p [=x(t)2] of the transmit signal x(t), and a distortion compensation coefficient storage unit 3c stores a distortion compensation coefficient h(p) conforming to each power of the transmit signal x(t) and outputs the distortion compensation coefficient hn−1(p) conforming to the power p of the transmit signal x(t). Further, the distortion compensating coefficient storage unit 3c updates the distortion compensation coefficient hn−1(p) by the distortion compensation coefficient hn(p) obtained by an LMS algorithm. A complex-conjugate signal output unit 3d outputs a complex-conjugate signal of a feedback demodulated signal y(t) that enters from the quadrature demodulator 14, and a subtractor 3e outputs the difference e(t) between the transmit signal x(t) and the feedback demodulated signal y(t). A multiplier 3f multiplies hn−1(p) and y*(t) together and outputs u*(t). A multiplier 3g multiplies e(t) and u*(t) together, a multiplier 3h multiplies the output signal of the multiplier 3g by a step-size parameter μ, and an adder 3i adds hn−1(p) and μe(t)u*t and inputs the resultant signal to the distortion compensating coefficient storage unit 3c. A calculation in accordance with the LMS algorithm set forth below is performed by the arrangement described above.hn(p)=hn−1(p)+μe(t)u*(t)  (1)e(t)=x(t)−y(t)y(t)=hn−1(p)x(t)f(p)u(t)=x(t)f(p)≈h*n−1(p)y(t)hn−1(p) h*n−1(p)≈1p=|x(t)|2where:                x(t): transmit signal        f(p): distortion function of transmit amplifier        h(p): estimated distortion compensation coefficient        μ: step-size parameter        y(t): feedback signal        u(t): distorted signal        
Further, x, y, f, h, u, e represent complex numbers and * signifies a complex conjugate. On the assumption that amplitude distortion of the amplifier is not very large [hn−1(p)·h*n−1(p)≈1], u(t) approximates h*n−1(p)·y(t). By executing the above-described processing, the distortion compensation coefficient h(p) is updated by Equation (1) in such a manner that the power of the difference e(t) between the transmit signal x(t) and the feedback signal y(t) will be minimized, and h(p) eventually converges to the optimum distortion compensation coefficient so that compensation is made for the distortion in the power amplifier 4.
With the arrangement of FIG. 5, a problem which arises is that when it is attempted to increase the number of carriers, there is a limitation upon the digital processing units, especially upon the sampling frequency Fs of the DA and AD conversions. For example, it order to double the number of carriers, it is necessary that the sampling frequency Fs of the digital units be doubled (Fs=12×W0). That is, since the sampling frequency Fs prior to doubling of the number of carriers is W0 in view of FIG. 6, Fs=12×W0 will hold if the number of carriers is doubled.
Thus, generally the sampling frequency of an AD converter represents a bottleneck. In a case where the frequency is not made higher, it is contemplated, as shown in FIG. 8, to split the carriers into two systems, namely systems on the low- and high-frequency sides, execute processing for transmitting a signal wirelessly in each system, and unite the processed signals through bandpass filters (BPF) 21, 22 downstream of transmit amplifiers 10, 10′. It should be noted that the radio communication apparatus shown in FIG. 8 is not well-known art.
In the multimedia-type radio communication apparatus of FIG. 8, the transmitting unit on the low-frequency side (the transmitting unit in the upper half of FIG. 8) is identical with that of the first example of the prior art shown in FIG. 5, and like components are designated by the same reference characters. Further, the transmitting unit on the high-frequency side (the transmitting unit in the lower half of FIG. 8) structurally is identical with that of the first example of the prior art shown in FIG. 5, and like components are designated by the same reference characters having “′” appended thereto. The sampling frequency Fs of the DA converters 6, 6′ and AD converters 13, 13′ is 6×W0, which is the same as the frequency prior to an increase in number of carriers. The transmitting unit on the high-frequency side differs from the transmitting unit on the low-frequency side in that whereas the local oscillator 9 on the low-frequency side generates a local signal of frequency f0, the local oscillator 9′ on the high-frequency side generates a local signal of frequency (f0+W0).
FIG. 9 is a diagram useful in describing the operation of the radio communication apparatus shown in FIG. 8. In FIG. 9, (1) to (8), (5)′ and (6)′ indicate spectrums associated with portions identified by the same numbers in FIG. 6. The spectrums at (1) to (7) are the same as those in FIG. 6, and the spectrums of (1) to (4) and (7) are not illustrated.
The local signal frequency that is output from the local oscillator 9 on the low-frequency side is f0, and the local signal frequency that is output from the local oscillator 9′ on the high-frequency side is (f0+W0). As a result, the band of the main signal on the high-frequency side is greater by W0 than the band of the main signal on the low-frequency side, as is evident from (5), (5)′ in FIG. 9. However, since the radio signal frequency is down-converted by (f0+W0) by the mixer 12′, the bands of the output signals of the AD converters 13, 13′ are the same, as evident from (6), (6)′ in FIG. 9.
The bandpass filters 21 and 22 have bandpass characteristics indicated by dashed lines A and B, respectively, in FIG. 9. As a result, the filters are capable of outputting radio signals that do not contain distortion D1, D2 and D1′, D2′.
Thus, in accordance with the example of the arrangement shown in FIG. 8, it is possible to cope even with a doubling of the number of carriers and, moreover, distortion outside the baseband region can be eliminated. With the above arrangement, however, it is required that the analog bandpass filters 21, 22 have a very steep attenuation characteristic in order to so arrange it that the signal from the low-frequency side will not flow into the high-frequency side and the signal from the high-frequency side will not flow into the low-frequency side. The problem which arises is that this is not realistic. Furthermore, eliminating the bandpass filters 21, 22 and relying upon direct coupling (wire coupling) is conceivable. If the bandpass filters 21, 22 are eliminated, however, the two signals will flow mutually into the transmit amplifiers 10, 10′ and third- and fifth-order intermodulation IM1 to IM4 will occur, as shown in FIG. 10.
Intermodulation distortion is a phenomenon in which when the signals of two waves enter a passive element such as an amplifier AMP, a distorted signal is produced where the frequency intervals of the two waves are separated, as illustrated at (A) in FIG. 10. If the intermodulation distortion is third-order intermodulation distortion, then it occurs at a frequency of f2+(f2−f1); if the intermodulation distortion is fifth-order intermodulation distortion, then it occurs at a frequency of f2+2 (f2−f1). If a modulated wave is considered to be a set of line spectrums, then, as a result of each line spectrum producing intermodulation distortion, distorted signals will be produced on both sides of the modulated wave, as indicated at (B) in FIG. 11, because a similar principle is at work.
Prior art for eliminating intermodulation produced when a plurality of signals have entered a transmit amplifier has been proposed in addition to the prior art of FIG. 5 and the arrangement shown in FIG. 8 (see the specification of JP 2002-64340A). This prior art, which has a loop for feeding the output signal of a power amplifier back to first and second distortion compensating units, feeds back low-frequency components of the output signal of the power amplifier to the first distortion compensating unit via a low-pass filter, thereby compensating for intermodulation distortion that occurs on the low-frequency side, and feeds back high-frequency components of the output signal of the power amplifier to the second distortion compensating unit via a high-pass filter, thereby compensating for intermodulation distortion that occurs on the high-frequency side.
The prior art of FIG. 5 is such that if an attempt is made to increase the number of carriers, a problem which arises is that there is a limitation upon the sampling frequency Fs of the DA and AD conversions. The number of carriers that can be added on is limited by this limitation on the sampling frequency.
The example of the arrangement of FIG. 8 is such that the analog bandpass filters require a very steep attenuation characteristic.
The prior art disclosed in the reference cited above eliminates intermodulation distortion but it is not art for increasing frequency. Further, this prior art does not eliminate intermodulation distortion produced when multicarriers are divided into two systems, namely systems on low- and high-frequency sides, and outputs of transmit amplifiers in transmitting units provided in respective ones of the systems are directly coupled.